Wide Dynamic Range Power Detection Scheme

ABSTRACT

Power detection system and method provide for a wide dynamic range power detection. The system may incorporate a coupler to sample a transmitted or received power. Such sampled power is then directed to a detector configured to generate a voltage from its rectified current. Such voltage after optional filtering and buffering is directed to multiple amplifiers wherein each amplifier has a distinct associated gain. For improved noise immunity, a differential output amplifier may be used. For low power levels, the output of an amplifier with a large gain is directed to an analog-to-digital converter (ADC), wherein the analog voltage is quantized into a digital value. As power is increased, the output of an amplifier stage with a smaller gain can then be directed to the ADC such that saturation of the ADC is avoided. Hysteresis is implemented so to avoid undesirable and unnecessary rapid switching.

REFERENCE TO EARLIER FILED APPLICATIONS

This application is a continuation of and hereby incorporates byreference U.S. patent application Ser. No. 10/443,157, on May 21, 2003,entitled “WIDE DYNAMIC R_sc ANGE POWER DETECTION SCHEME.”

INTRODUCTION

1. Field

The present invention relates generally to the field of electroniccommunications. More particularly, the present invention relates to thepower measurement and control of a signal transmitted over a medium.

2. Background

In developing a communications system, it is generally advantageous fora communications link to utilize the strongest signal feasible forimproving signal quality and for providing sufficient coverage or range.With regard to signal quality, a stronger signal yields a highersignal-to-noise ratio (SNR). Also, a stronger signal propagates afurther distance. However, the power level of transmitted signals mustbe constrained within limits. For example, in most situations,transmission power levels are regulated under rules imposed bygovernmental agencies such as the Federal Communications Commission(FCC). Indeed, this is important so as to prevent one or more powerfulsignals from interfering with the communications of other signals in thesame frequency range. Other restrictions may be imposed by standardscommittees or may be self-imposed by a system in order to minimizeinterference where several signals are expected to exist simultaneously.

In view of the above, an important consideration in designing acommunication system is its performance over a wide temperature range.It has been observed that the characteristics of a communication systemchange over temperature in such a way that its transmission power isaffected. For example, while maintaining all other conditions constant,a communication system can transmit lower power levels at elevatedtemperatures and it can transmit higher power levels at very coldtemperatures, and vice-versa.

Whatever the characteristics of a communications system may be, it isnonetheless desirable to closely monitor and control the transmissionpower. It is therefore important to know the transmission power levelsof communication system. A common scheme for monitoring or detecting asystem's radio frequency (RF) power is through the use of asemiconductor Schottky barrier diode. RF detectors are essentially lowsensitivity receivers which function on the basis of directrectification of an RF signal through the use of a non-linear resistiveelement—a diode. Generally detectors using Schottky diodes can beclassified into two distinct types: the small-signal type, also known assquare-law detectors; and the large-signal type, also known as linear orpeak detectors. In operation, a small-signal detector is dependent onthe slope and curvature of the VI characteristic of the diode in theneighborhood of the bias point. The output of the detector isproportional to the power input to the diode. That is, the outputvoltage (or current) of the detector is proportional to the square ofthe input voltage (or current), hence the term “square law.”Large-signal detector operation is dependent on the slope of the VIcharacteristics in the linear portion, where the diode functionsessentially as a switch. In large-signal detection, the diode conductsover a portion of the input cycle and the output current of the diodefollows the peaks of the input signal waveform with a linearrelationship between the output current and the input voltage.

The square law dynamic range may be defined as the difference betweenthe power input for a 1 dB deviation from the ideal square law response(compression point) and the power input corresponding to the tangentialsignal sensitivity (TSS). TSS is a measure of low level sensitivity withrespect to noise. Normal operating conditions for the Schottky detector,a square-law detector, call for a large load resistance (100 kΩ) and asmall bias current (20 μA). These normal conditions assure the minimumvalue of TTS, but not the maximum value of compression level.

One conventional manner of raiding the compression level is by reducingthe value of the load resistance, RL. But the sensitivity degrades bythe factor RL/(RL+RV), where RV is the diode's specified videoresistance. The degradation in TSS can exceed the improvement incompression, such that there is no improvement in square lay dynamicrange. Another conventional technique for raising the compression levelis to increase the bias current. This also degrades the sensitivity, butthe improvement in compression exceeds this degradation so square lawdynamic range is increased. Although these approaches may improve adetector's general performance, they do not significantly improve adetector's dynamic range.

With regard to a communication system, it is important to know itstransmission power level at particular temperatures of operation.Conventional approaches have used a power detector and a temperaturesensor so as to develop calibration table. In conventional calibrationmethods, the entire communication system is exercised at varioustemperatures while noting the output of the detector module. When placedin service, the communication system would then retrieve calibrationdata at a measured temperature so as to accurately measure the system'stransmission power. Such conventional calibration methods, however,necessarily required that the entire system, or at least a large part ofit, be placed in a temperature chamber. Because of the size and mass ofsuch configurations, the calibration system is slow. Moreover, becausean entire system is calibrated, and changes in components, such as upona failure, required re-calibration.

SUMMARY

The present invention as embodied and broadly described herein providesadvantages over conventional methods by providing accurate measurementsof a communication system's transmission power over an extended dynamicrange. In one embodiment, a method for detecting power of acommunication system may comprise: providing a source signal that has apower level within a range; generating a detector signal with amagnitude relative to power level of the source signal; generating aplurality if amplified detector signals each of which being a product ofamplification by a gain of the detector signal, each gain being distinctand having a predetermined gain value; selecting as an output signal oneof the plurality of amplified detector signals; switching the selectionas output signal to another one of the plurality of amplified detectorsignals in response to the output signal reaching one of a plurality ofthresholds, wherein the plurality of thresholds are predetermined, in apre-calibration, based on the predetermined gain values and amplitudesof the output signal in order to substantially avoid saturation andsense detectable power levels across the range; and associating thepower level to the output signal, wherein the switching increases therange of power levels that are detectable.

In another possible embodiment, a system for detecting power in acommunication system may include: means for providing a source signalthat has a power level within a range; means for generation a detectorsignal with a magnitude relative to the power level of the sourcesignal; means for generating a plurality of amplified detector signalseach of which being a product of amplification by a gain of the detectorsignal, each gain being distinct and having a predetermined gain value;means for selecting as an output signal one of the plurality ofamplified detector signals; means for switching the selection as outputsignal to another one of the plurality of amplified detector signals inresponse to the output signal reaching one of a plurality of thresholds,wherein the plurality of thresholds are predetermined, in apre-calibration, based on the predetermined gain values and amplitudesof the output signal in order to substantially avoid saturation andsense detectable power levels across the range; and means forassociating the power level to the output signal, wherein the switchingincreases the range of power levels that are detectable.

The method may include coupling a portion of the source signal toproduce the detector signal. The method may also include filteringand/or buffering the detector signal, for example, prior to amplifyingthe detector signal. The source signal may be a radio frequency (RF)signal.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings, which are incorporated in and form a part ofthis specification, illustrate embodiments and, together with thedescription, serve to explain the principles of the present disclosure.

FIG. 1 is a schematic diagram of a configuration for measuring a powersignal.

FIG. 2 is a schematic diagram of another configuration for measuring apower signal.

FIG. 3 is a graph of response curves for a power detection circuit.

FIG. 4 is a graph of response curves for a power detection circuit.

FIG. 5 is a block diagram of a traditional approach for measuring theoutput power of a communication system.

FIG. 6 is a block diagram of a representative embodiment of the newapproach to measuring the output power a communication system.

FIG. 7 is a mechanical drawing of a front, side and rear view of adetector module.

DETAILED DESCRIPTION

The present invention provides a new transceiver power detectionarchitecture. A representative embodiment of the new architecture callsfor placement of a detector module at or near the transceiveroutput/input (e.g., at or near the antenna connection point). Moreover,the present invention provides a new detector module that provides awide power detection range.

According to an aspect of the invention, a power detection circuitincorporates a coupler to sample a transmitted or received power. Suchsampled power is then directed to a detector diode configured togenerate a voltage from its rectified current. Such voltage afteroptional filtering and buffering is then directed to multiple amplifierswherein each amplifier has a distinct associated gain. For improvednoise immunity, a differential output amplifier may be used. For lowpower levels, the output of an amplifier with a large gain is directedto an analog-to-digital converter (ADC), wherein the analog voltage isquantized into a digital value. As power is increased, the output of anamplifier stage with a smaller gain can then be directed (e.g.,switched) to the ADC such that saturation of the ADC is avoided. Acircuit with two amplifiers will be described, however, the presentinvention is not so limited. In fact, more stages can be implemented toextend the dynamic range of the detection circuit over a wider range.

According to another aspect of the invention, a power detector modulewith a small associated thermal mass is advantageously placed near atransceiver's output, such as an antenna, so as to provide accuratemeasurements of the transceiver's output power. Moreover, the powerdetector module is calibrated over a wide range of temperatures suchthat a power detector module reading at a particular temperature can beaccurately associated with a calibrated output power. In response to anaccurate measurement of the output power, an input signal of thetransceiver can be attenuated or amplified, so as to closely control itsoutput power. In the discussion to follow attenuation or amplificationwill generally be referred to as amplification where it is understoodthat amplification can produce a gain equal to or greater that one(i.e., A≧1) or a gain equal to or less than one (i.e., A≦1), alsoreferred to as attenuation; moreover, buffering is achieved with a gainequal to one (i.e., A=1). Where amplification is mentioned, gain orattenuation is also appropriate. Although the discussion focuses for themost part of the transmitter portion, the power measurement at thisphysical location applies also to the input of the receiver portion ofsuch transceiver.

In the description to follow, certain aspects of the present inventionare described in detail. So as not to complicate the discussionpresented here with unnecessary detail, certain aspects have beensimplified. Nevertheless, because the present invention finds generalapplication in many systems, these simplifications do not imply narrowapplicability.

Shown in FIG. 1 is a schematic diagram of a circuit 100 used to measurethe transmission power of a transmitter. As shown, radio frequency (RF)source 102 provides a signal having a power level RF output 104. RFoutput 104 can take various forms including, for example, an antenna. Atransmission medium 103 is used to direct the signal from RF source 102to RF output 104. Transmission medium 103 can take various formsincluding, for example, a waveguide or coaxial conductor.

The traditional approach for measuring the power of the signal emanatingfrom RF output 104 has been to sample a small proportion of the power atRF source 102 through the use of coupler 106. RF coupler 106 caninclude, for example, a microstrip coupler. Coupler 106 therebyredirects a small portion of the signal to be processed by detectioncircuitry. It is preferred that coupler 106 redirect only a very smallportion of the signal. Coupler 106 directs its sampled signal todetector diode 108.

Detector diode 108 can be, for example, a semiconductor device such as aSchottky barrier diode configured to rectify an RF signal. In operation,detector diode 108 functions as a small signal detector within itsconduction region. For small currents, it may be necessary to biasdetector diode 108 to place it in its conduction region. Detector bias110 provides such a function. When the sampled signal is sufficientlylarge, less biasing of detector diode 108 may be necessary. Accordingly,detector bias control signal 112 may be used to properly control theamount of biasing.

Resistor 108 in conjunction with the current rectified by detector diode108 produces a voltage that is then input to amplifier 114. Amplifier114 has an associated and predetermined gain, G1, and is furtherconfigured to produce a differential output from a single-ended input.Power detect+signal 116 and power detect−signal 118 provide such adifferential output. In a calibration process, power detect +/−signals116/118 can be associated with known RF output 104 power levels.Thereafter, measured power detect +/−signals 116/118 can be used todetermine an RF output 104 power level. A disadvantage of conventionalcircuit 100 is that it has a relatively narrow dynamic range that canonly be extended minimally through manipulation of the detector bias.When configured to measure relatively large power levels, conventionalcircuit 100 may quickly become saturated. Moreover, conventional circuit100 experiences difficulty in detecting low power levels whereby changesin voltage are of the magnitude of the quantization error of anassociated ADC (see discussion of FIG. 6 below).

Shown in FIG. 2 is a schematic diagram of a wide dynamic range circuit200 used to measure the transmission power of a transmitter according tothe representative embodiment of the present invention. As shown, RFsource 202 provides a signal having a power level to RF output 204. RFoutput 204 can take various forms including, for example, an antenna. Atransmission medium 203 is used to direct the signal from RF source 202to RF output 204. Transmission medium 203 can take various formsincluding, for example, a waveguide or coaxial conductor.

The power of the signal emanating from RF output 204 could have beenmeasured by sampling the power at RF source 202 with the coupler 206positioned at the RF source. But because the signal can degrade in thepath from source 202 to RF output 204, the power is sampled in thisimplementation at or near RF output 204 as shown by the proximity ofcoupler 206 to RF output 204. RF coupler 206 can include, for example, amicrostrip coupler. Coupler 206 thereby redirects a small portion of thesignal at or near RF output 204 to be processed by detection circuitryaccording to the present invention. It is preferred that coupler 206redirect only a small portion of the signal. Coupler 206 directs itssampled signal to RF amplifier 208 where the sampled signal is amplifiedto a nominal signal that is directed to detector diode 212.

Amplifier bias 210 is configured to set the appropriate gain for RFamplifier 208 such that the output of RF amplifier 208 is at a nominallevel. Detector diode 212 can be, for example, a semiconductor Schottkydevice configured to rectify an RF signal. In this embodiment, Schottkydiode detectors are used to detect small signals close to the noiselevel and to monitor large signals well above the noise level. For thenoise level up to about −20 dBm the slope of the response curves isclose to constant. This is the square law region. The diode receives thesignal directly from an antenna in most systems, although a preamplifiersuch as RF amplifier 208 is typically used to improve sensitivity. Thistype of receiver, appropriate for RF applications, is used in shortrange radar or in counter-measure equipment where the sensitivity of amore complicated superheterodyne receiver is not needed. Above about −10dBm the slope is closer to linear but may vary about 30% for differentvalues of frequency, diode capacitance, and load resistance. The slopemay be controlled by tuning at the proper power level.

Linear detection is used in power monitors. In some applications thelinearity is important because the detected voltage is a measure ofpower input. In other applications, linearity is not as importantbecause lookup tables are developed that associate a particular powerlevel with a particular diode current or voltage, however,discernability is necessary.

Over a wide range of input, power level, P, is monotonically related tothe output voltage, V. At low levels, below −20 dBm, a detector diodeoperates in its square law region. When DC bias current (usuallymicroamperes) is used, the diode impedance is independent of power levelsuch that tuning can be done at any level. Typically, a diode is tunedat −30 dBm. The detected voltage at this level is called the voltagesensitivity. At higher power levels the diode impedance changes withpower. The slope is largely related to diode capacitance, frequency, andload resistance.

As previously noted, in operation, detector diode 212 is in itsconduction region. For small currents, it may therefore be necessary tobias detector diode 212 to place it in its conduction region. Detectorbias 214 provides such a function. When the sampled signal issufficiently large, less biasing of detector diode 212 may be necessary.Accordingly, detector bias control signal 240 may be used to properlycontrol the amount of biasing. Resistor 213 in conjunction with thecurrent conducted by detector diode 212 produces a voltage that ispassed through low pass filter 216 and then to amplifier 218.

Amplifier 218 can be set for unity gain so as to act as a high inputimpedance buffer. Indeed, any gain setting can provide a bufferingeffect. The buffered signal is then directed to amplifiers 205 and 226having respective gains G1 and G2, wherein G1 is greater than G2. Withregard to amplifier 218, it provides isolation of amplifiers 205 and 226from the detector diode and further provides increased fanout withoutaffecting the detector circuitry. In fact, an embodiment can beimplements by incorporating further amplifiers having different gains.

Amplifier 205 has an associated and predetermined gain, G1, and isfurther configured to produce differential outputs from a signal-endedinput. Power detect+signal 222 and power detect−signal 224 provide suchdifferential outputs. In a calibration process, power detect +/−signals222/224 can be associated with known RF output 204 power levels.Thereafter, measured levels of power detect +/−signals 222/224 are usedto determine the corresponding RF output 204 power levels. Amplifier 205is similarly configured as amplifier 226, but with gain G2 anddifferential +/−outputs 228/230. It is noted that G1 is a relativelylarge gain as compared to G2, and the different gains address differentconduction regions of the detector, thereby producing the wide rangegain as will be further explained below.

Circuit 200 is further configured with temperature sensor 232 to producetemperature signal 234. Accordingly, in a calibration process such asthat described below, power detect +/−signals 222/224 and power detect+/−signals 228/230 as well as temperature signal 234 can be associatedwith known RF output 204 power levels at known temperatures. Thereafter,measured power detect +/−signals 222/224 and 228/230 and measuredtemperature signals 234 can be used to determined an RF output 204 powerlevel.

Circuit 200 as shown in FIG. 2 can be built as a single module orcircuit board. Moreover, circuit 200 can be incorporated into systemssuch as communication system 600 of FIG. 6 to be discussed below. Indoing so, connector 244 provides a convenient interface to othercircuitry. Connector 244 can thereby couple power detect +/−signals222/224 and 228/230, temperature signal 234, detector bias control 240,as well as power supply +5/−5 volt inputs 236/238 are ground 242. Powersupply +5/−5 volt provide the necessary power to the circuitry ofcircuit 200. Ground 242 provides a common reference to the circuitry ofthe circuit 200 as well as circuitry external to circuit 200.

The operation of circuit 200 can further be understood with reference tothe curves of FIG. 3. As shown, curve 304 corresponds to the output ofamplifier 205 that has a gain G1 greater than that of amplifier 226 thathas a gain of G2. For very small currents through detector diode 212, arelatively large gain such as G1 provides adequate change in voltage fora given change in power. For example, at a power level P1 a voltage V1is generated by amplifier 205 such that an increase from a power levelP1 to an incrementally larger P1+ΔP produces an incrementally largervoltage V1+ΔV.

In increasing the power from power level P1, the voltage follows thecurve 304 from V1. For example, at power level P2, amplifier 205produces a voltage V2; at power level P3, it produces a voltage of V4;at power level P4, it produces a voltage V5. At this point, it is notedthat the output voltage of amplifier 205 is approaching a maximumdetectable voltage, V6 (corresponding to power level P5), of ananalog-to-digital converter. Accordingly, it is undesirable to provide avoltage greater than V6, however, it may be desirable to measure powerlevels greater than P5 that would exceed the associated maximum voltageV6 using the gain setting of amplifier 205.

In order to prevent saturating ADC 256, among other things, circuit 200switches over from the output of amplifier 205 to the output ofamplifier 226, wherein amplifier 226 has a gain G2 that is less than theG1 of amplifier 205. Switching from amplifier 205 to amplifier 226 canbe achieved in many ways, including though the use of a multiplexer orsemiconductor switch 250 shown in FIG. 2. The appropriate signals arethen directed to ADC 256; its output 258 is directed to amicroprocessor. The transition from amplifier 205 to amplifier 226preferably occurs at an associated voltage V5 that is less than themaximum voltage V6. Accordingly, the power level P4 has an associatedvoltage V5 from amplifier 205, but the same power level P4 has a lowerassociated voltage V3 from amplifier 226.

For power level greater than P4, amplifier 226 provides associatedoutput voltages up to the maximum detectable power level P6 that isassociated with the maximum voltage V6 along curve 302. When amplifier226 is being used (i.e., curve 302 is being traversed), operation toamplifier 205 is not switched at power level, P4, but rather is switchedat power level P3. In this way, hysteresis 314 is produced andunnecessary and frequent switching can be avoided.

Note that for power levels less than P3 along curve 302, incrementalchanges in power produce small changes in voltage such that differencein power level may not be discernable by an ADC. In certain situations,the changes in voltage are at the level of quantization of the ADC. Insuch situations, detector diode 212 may be biased more aggressively bymeans of detector bias 214 and detector bias control 240. Moreover, RFamplifier 208 may be biased more aggressively by means of amplifier bias210. More conveniently, operation may be switched from amplifier 226 toamplifier 205 such that curve 304 is traversed for power levels belowP3.

While traversing curve 304, reduced sensitivity may nonetheless beexperienced for low power levels below P1, for example, such thatfurther amplifier stages with larger gains can be implemented. FIG. 4shows the response curves of various outputs associated with amplifierswith different gains. Note that curves 402 has been added that isassociated with an amplifier stage that has an associated gain greaterthan G1 of amplifier 205. Accordingly, power levels below P1 down to P7can be detected. Traversal of the various curves 402, 302 and 304 istherefore controlled by switching amongst the various amplifiers. Asbefore hysteresis such as hysteresis 404 is desirable to avoidunnecessary and frequent switching amongst amplifiers.

In another embodiment, the present invention addresses issuessurrounding control of the output power of a radio transmitter (andmonitoring of input power at the input of the receiver). Indeed, becauseof very tight government regulations and because of very transmitter beclosely controlled.

To better understand this aspect of the present invention, however, anunderstanding of traditional approaches provides a useful context. Atraditional approach toward measuring and, in turn, controllingtransmitter output power is as shown in communication system 500 of theblock diagram FIG. 5. As shown, communication system 500 comprises atransmitter 502 and receiver 504. Transmitter 502 is generallyconfigured to modulate and condition signals for transmission over amedium (e.g., wireless medium) such as by means of antenna 526.Conversely, receiver 504 is generally configured to demodulate a signalreceived over a medium such as by means of antenna 526.

With regard to the transmission aspects of communication system 500,amplifier 506 is configured to receive an input signal. In certainimplementations, amplifier 506 can be a variable attenuator or avariable gain amplifier. In communication system 500, the input signalgenerally contains information such as digital information in amodulated form (e.g., QAM modulated signal) or analog information with aspecified bandwidth. The output of amplifier 506 is coupled to mixer508. In conjunction with synthesizer 510, mixer 508 is used toup-convert the input signal to another frequency usually a higherfrequency appropriate for a transmission medium being used, such as anRF for wireless communication. The amplifier 512 amplifies the RF signalto a higher power appropriate for transmission over a medium. Amplifier512 can be in many forms including integrated circuit amplifiers,magnetrons or traveling wave tubes (TWT).

Detector module 514, containing detector diode 516, is configured tomeasure a detected signal level at the output of amplifier 512 withintransmitter 502. This detected signal level is used to approximate theoutput power transmitted from antenna 526. Importantly, transmitter 502can be a large module with many components and significant thermal mass.Signal converter 518 is provided to condition detected signals forcoupling them to microprocessor 520 which in turn controls the level ofamplification of amplifier 506. Signal converter 518 include detectorbiasing and amplification as well as filtering. By varying theamplification of amplifier 506 according to the signal detected bydetector module 514, the have used microstrip proximity couplers fordirecting a detected signal to detector diode 516.

Signal isolator 522 is provided between amplifier 512 and diplexer 524to minimize return losses within communication system 500. Note that incertain implementations a circulator is used in place of signal isolator522. Whichever one is used, signal isolation is achieved to provide veryclosely matched signals. Diplexer 524 allows for dual transmit andreceive functions within communication system 500. In certainapplications, diplexer 524 is a bandpass filter that separatestransmitted and received signals existing within different frequencyranges. Accordingly, an amplified RF signal generated by amplifier 512is directed through diplexer 524 and is passed to antenna 526 fortransmission over a medium, a wireless medium in this example.Conversely, where a signal is received by antenna 526, such receivedsignal is directed through diplexer 524 and passed to receiver 504.Because details of receiver 504 are not necessary to understand thepresent invention, receiver 504 will not be further described.

The traditional approach of FIG. 5 can further be understood as acontrol system problem wherein it is desired to control the output powerof a signal exiting antenna 526 by controlling an input signal toamplifier 506. For proper control of communication system 500,observability and controllability issues must be considered. Asdescribed, the output power of a signal exiting antenna 526 isobservable by monitoring the power of a signal that exists at detectormodule 514. However, there are intervening complications. Here we notethat the output power is not observed directly, but rather in a detachedand indirect way. As shown, isolator 522 and diplexer 524, with alltheir real-world complexities, reside between the desired signal to beobserved (the output power) and the signal actually measured (the signalat detector module 514). Thus, the signal observed at detector module514 is not a true observation of the output signal from antenna 526 suchthat it is necessary to account for the characteristics of more thanjust detector module 514 at various temperatures and frequencies overtime. For example, as described, traditional approaches characterize theentire communication system 500 or at least transmitter 502 over atemperature range.

In controlling the output power of a communication system, it is furtherimportant to consider the effects of temperature where communicationsystem 500, including transmitter 502, may be exposed to widetemperature ranges. Components of communication system 500 and inparticular transmitter 502, signal isolator 522, and diplexer 524 canexhibit performance changes as a function of temperature. For example,transmitter circuitry such as that contained within mixer 508,synthesizer 510 and amplifier 512 may vary in unknown or unpredictableways as a function of temperature. This is further exacerbated by thefact that detector module 514 (including detector diode 516) and signalconverter 518 also experience changes as a function of temperature.

In order to account for changes over temperature, traditional approacheshave performed temperature calibration of communication system 500 or atleast transmitter 502 by enclosing them in a temperature chamber andobtaining calibration data by precisely measuring an output power levelfrom antenna 526 at specified temperatures. In this way calibration dataover a known temperature range has been associated with a transmittedoutput power. In operation, temperature sensor 528 would provide atemperature signal to mircoprocessor 520. Temperature sensor 528 isimplemented in many forms including a temperature sensing diode orresistor. Microprocessor 520, through the use of calibration dataprovided in a lookup table, for example, would then convert a receiveddetector signal to a calibrated power level. An appropriate signal wouldthen be directed by mircoprocessor 520 to amplifier 506 which willcontrol the output power to communication system 500 in a known way.

As described, the traditional approach to calibrating a communicationsystem requires that the entire communication system or at leasttransmitter 502 be placed within a temperature chamber for calibration.Because of the thermal mass associated with even just transmitter 502,temperature calibration was a lengthy process wherein it was necessarythat components be left at a particular temperature for an extendedperiod of time so as to assure that all the components were at thespecified temperature. In obtaining calibration data with fineresolution over a wide temperature range, the time and cost ofcalibration increases dramatically. Moreover, because communicationsystem 500 or at least 502 can be large, a temperature chamber islimited in the number of units it can calibrate at a time.

With this understanding of traditional methods of calibratingcommunication systems, the following embodiment of the present inventioncan be better appreciated. FIG. 6 is a diagram of communication system600 according to a representative embodiment of the present invention.System 600 includes receiver 604, diplexer 624 and antenna 626 much likecommunication system 500 of FIG. 5. But communication system 600 differssignificantly from traditional approaches in the placement of detectormodule 614 that includes detector diode 616, signal converter 618, andtemperature sensor 628.

In this embodiment, detector module 614 is placed on a substrate thatfurther includes a waveguide coupled between the path from diplexer 624to antenna 626, wherein diplexer 624 is used to switch between thetransmission and receiving modes of communication system 600. Othertypes of coupling for detecting a power level are also appropriate. Theposition of detector module 614 close to antenna 626 allows anyvariations in the transmitter path (e.g., transmitter components,isolator, circulator, or diplexer filter) without degrading thecalibrated power accuracy of the detector as the transmitter path variesover frequency, power, or time. In addition, the physically smallerdetector module 614 allows for faster thermal cycling, improvingcalibration time and easing manufacturability of communications system600.

Signal converter 618 is also provided which includes ADC that convertsthe analog signal from detector diode 616 into a digital signalappropriate for input to microprocessor 620. In order to do so, an ADCdetects a voltage and quantizes such voltage into a digital value. Theresolution of such an ADC is limited by its order. For example an 8-bitADC that can quantize a signal over 5-volt range divides the 5-voltrange into 28=256 equally spaced voltage gradation whereas a 12-bit ADCdivides 5- volt range into 212=4096 equally spaced gradations. Notsurprisingly, a higher order ADC has a higher associated cost and can bemore complex in its integration.

In seeking to obtain accurate measurements of the output power ofcommunication system 600 over a temperature range, the present inventiondoes not require placing a large thermal mass, such as that ofcommunication system 600 or transmitter 602, in a temperature chamber.With the advantageous placement of detector module 614 near the outputof communication system 600, it is not necessary to calibrate theoperation of the entire system. Rather, it is only necessary tocalibrate detector module 614 with its associated smaller thermal mass.Indeed, it is not important that the characteristics of transmitter 602,isolator 622, or diplexer 624 change, and all that matters forcalibration is that the power level of the output signal is known. Thiscalibration can be done by passing a signal of known power level throughdetector module 614 over a temperature range.

The design of communication system 600 can be further understood interms of a control system problem wherein it is desired to control theoutput power of a signal exiting antenna 626 by controlling an inputsignal to amplifier 606. Observability and controllability issues musttherefore be considered. With reference to communication system 600 ofFIG. 6, however, the output power of a signal exiting antenna 626, whichdoes not vary significantly over temperature, is well observed throughmonitoring of the power of a signal immediately before antenna 626.Advantageously, with this architecture, there are minimal interveningcomplications. Here, the output power is observed almost directlyinstead of in an indirect way. As shown in FIG. 6, transmitter 602,isolator 622 and diplexer 624, with all their real-world complexities,reside before detector module 614. Thus, the signal observed at detectormodule 614 is a good observation of the output signal at antenna 626,and the only accounting needed is an accounting for the characteristicsof detector module 614 over a temperature range.

It is important to note that the power of the signal measured bydetector module 614 also includes a contribution from a signal receivesby antenna 626. This received signal, however, is many orders ofmagnitude lower than the transmitted signal such that any errorintroduced by this signal is insignificant. For example, wherecommunication system 600 may be transmitting at 30 dBm, it may bereceiving at −60 dBm which is many orders of magnitude lower. It isfurther important to note that in a preferred embodiment, antenna 626 isclosely matched to the rest of the system such that return losses arevery low and, in turn, do not contribute in a significant amount to themeasurement of detector module 614.

In light of the aforementioned complications of traditional calibrationmethods, it is therefore notable that, for calibration, the presentinvention requires placement in a temperature chamber only of thedetector module. In this way, a plurality of detector modules can becalibrated within a temperature chamber at once. Moreover, becausedetector module 614 has a smaller thermal mass than communication system600 or transmitter 602, detector module 614 can reach a desiredtemperature much faster, thereby decreasing calibration time andimproving manufacturability while reducing associated costs.

In a calibration process, detector module 614 is tested at varioustemperatures by passing a signal of known frequency and power levelthrough detector module 614 and measuring the output of signal converter618. A lookup table is generated that associate signal levels fromdetector module 614 to known signal power levels of known frequency.After calibration, such a lookup table can then be stored inmicroprocessor 620 or an appropriate memory (not shown). In operation,communication system 600 and its various components including detectormodule 614 may operate over a wide range of temperatures. At a giventemperature, however, as measured by temperature sensor 628 withindetector module 614, the power level of a signal passing from diplexer624 to antenna 626 is determined by the microprocessor 620 through theuse of the lookup table. In turn, microprocessor 620 directs a signal toamplifier 606 to appropriately amplify the input signal level which thenaffects the output power level and the power level measured by detectormodule 614. A feedback loop is therefore provided through which theoutput power level of communication system 600 is closely controlled byknowing, in a calibrated way, the characteristics of detector module614.

Shown in FIG. 7 is a mechanical drawing with front, side, and rearperspective of a detector module 700 according to the present invention.The dimension of waveguide 702 are provided as appropriate for signalsof the required frequency. Waveguide 702 is made of conductive material.A signal such as an RF signal is coupled to detector probe 704 that ispositioned to protrude into the waveguide 702 opening. Detector probe704 is further coupled to detector circuitry 706, signal convertercircuitry 708, and temperature sensor 712 which are contained withindetector module 700. Although waveguide 702 is made of a conductivematerial, detector module 700 need not be a conductive material.Connector 710 provides an interface between the circuitry containedwithin detector module 700 and other components of a communicationsystem such as a microprocessor or a transmitter (not shown). From FIG.7, it can therefore be appreciated that the small size of detectormodule 700 facilitates calibration within a temperature chamber.

Described with reference to FIG. 6 was detector diode 616 and certaingenerally described circuitry. Whereas the system discussed above may beimplemented with conventional detector diodes and circuitry, anotheraspect of the present invention incorporates improved circuitry thatwidens the dynamic operating range of the detector diode as describedabove.

While various embodiments and advantages have been described, it will berecognized that a number of variations will be readily apparent. Thus,the present teachings may be widely applied consistent with theforegoing disclosure and the claims which follow.

1. A method for detecting power in a communication system, comprising:providing a source signal that has a power level within a range;generating a detector signal with a magnitude relative to the powerlevel of the source signal; generating a plurality of amplified detectorsignals each of which being a product of amplification by a gain of thedetector signal, each gain being distinct and having a predeterminedgain value; selecting as an output signal one of the plurality ofamplified detector signals; switching the selection as output signal toanother one of the plurality of amplified detector signals in responseto the output signal reaching one of a plurality of thresholds, whereinthe plurality of thresholds are predetermined, in a pre-calibration,based on the predetermined gain values and amplitudes of the outputsignal in order to substantially avoid saturation and sense detectablepower levels across the range; and associating the power level to theoutput signal, wherein the switching increases the range of power levelsthat are detectable.
 2. The method of claim 1, wherein the switching toanother one of the plurality of amplified detector signals substantiallyavoids saturation by switching to a lower predetermined gain value foramplification of the detector signal.
 3. The method of claim 2, whereinthe switching to another one of the plurality of amplified detectorsignals allows sensing detectable power levels substantially across therange by switching to a different predetermined gain value foramplification of the detector signal.
 4. The method of claim 3, whereinthe different predetermined gain value for amplification of the detectorsignal is either higher or lower than the predetermined gain valuepreceding it, depending on the output signal approaching amplitude ofsaturation or undetectably small power level.
 5. The method of claim 1,wherein first and second thresholds from among the plurality ofthresholds are associated with first and second predetermined gainvalues, respectively, and wherein the first threshold is higher than thesecond threshold and the first predetermined gain value is higher thanthe second predetermined gain value.
 6. The method of claim 5, whereinthe first predetermined gain value is used to produce the selected oneof the amplified detector signals and the second predetermined gainvalues is used to produce the selected other one of the amplifieddetector signals, such that the first threshold determines whenswitching from one to another one of the amplified detector signals. 7.The method of claim 1, wherein the power level of the source signal isdetected via the association to the output signal.
 8. The method ofclaim 1, further comprising coupling a portion of the source signal toan input of a detector by sampling signal transmission over a mediumbetween an RF (radio frequency) signal source and an antenna.
 9. Themethod of claim 1, further comprising filtering the detector signal. 10.The method of claim 1, further comprising buffering the detector signal.11. The method of claim 8, further comprising basing the detector. 12.The method of claim 1, wherein the switching includes hysteresis tosubstantially avoid rapid switching.
 13. A system for detecting power ina communication system, comprising: means for providing a source signalthat has a power level within a range; means for generating a detectorsignal with a magnitude relative to the power level of the sourcesignal; means for generating a plurality of amplified detector signalseach of which being a product of amplification by a gain of the detectorsignal, each gain being distinct and having a predetermined gain value;means for selecting as an output signal one of the plurality ofamplified detector signals; means for switching the selection as outputsignal to another one of the plurality of amplified detector signals inresponse to the output signal reaching one of a plurality of thresholds,wherein the plurality of thresholds are predetermined, in apre-calibration, based on the predetermined gain values and amplitudesof the output signal in order to substantially avoid saturation andsense detectable power levels across the range; and means forassociating the power level to the output signal, wherein the switchingincreases the range of power levels that are detectable.
 14. The systemof claim 13, wherein the switching to another one of the plurality ofamplified detector signals substantially avoids saturation by switchingto a lower predetermined gain value for amplification of the detectorsignal.
 15. The system of claim 13, wherein the switching to another oneof the plurality of amplified detector signals allows sensing detectablepower levels substantially across the range by switching to a differentpredetermined gain value for amplification of the detector signal. 16.The system of claim 15, wherein the different predetermined gain valuefor amplification of the detector signal is either higher or lower thanthe predetermined gain value preceding it, depending on the outputsignal approaching amplitude of saturation or undetectable small powerlevel.
 17. The system of claim 13, wherein first and second thresholdsfrom among the plurality of thresholds are associated with first andsecond predetermined gain values, respectively, and wherein the firstthreshold is higher than the second threshold and the firstpredetermined gain value is higher than the second predetermined gainvalue.
 18. The system of claim 17, wherein the first predetermined gainvalue is used to produce the selected one of the amplified detectorsignals and the second predetermined gain values is used to produce theselected other one of the amplified detector signals, such that thefirst threshold determines when switching from one to another one of theamplified detector signals.
 19. The system of claim 13, furthercomprising signal conversion means for receiving the output signal andproducing a signal for used in the association of the output signal topower level of the source signal.
 20. The system of claim 19, whereinthe signal conversion means includes an analog-to-digital (ADC)converter.
 21. The system of claim 13, wherein the means for generatingthe detector signal includes a detector, the system further comprisingmeans for coupling a portion of the source signal to an input of thedetector by sampling source signal transmission over a medium between anRF (radio frequency) signal source and an antenna.
 22. The system ofclaim 13, further comprising means for filtering the detector signal.23. The system of claim 13, further comprising means for buffering thedetector signal.
 24. The system of claim 18, further comprising meansfor basing the detector.
 25. The system of claim 13, wherein the meansfor switching provides hysteresis to substantially avoid rapidswitching.